1. Field of the Invention
The present invention relates to radio positioning systems generally, and more particularly to improved methods of finding the positions of mobile terminals in radio communication systems, especially those employing Code Division Multiple Access (CDMA) technology.
2. Description of the Related Art
There are many systems known by which the position of a mobile terminal operating in a radio communications network may be determined. These include using the signals from transmitters not connected with the network, such as the Global Positioning System (GPS) satellites, but others make use of the signals radiated by the mobile terminal and picked up by remote receivers, such as the Time Of Arrival (TOA) and so-called “Radio Finger Printing” systems or, vice versa, using the signals radiated by the network itself and picked up by the mobile terminal. Chief amongst the last category are the Enhanced Observed Time Difference (E-OTD) and Observed Time Difference Of Arrival (OTDOA) systems.
The E-OTD system, although generally applicable to many different communication technologies, has been particularly applied to the Global System for Mobiles (GSM). Two principal, and different, methods of using the timing offsets of signals received from the network transmitters in the position computation have been described in the art. In one, e.g. EP-A-0767594, WO-A-9730360, U.S. Pat. No. 6,108,553 and AU-B-716647, the signals measured by a fixed receiver are used, in effect, to ‘synchronise’ the transmissions from the different transmitters. The instantaneous transmission time offsets of each transmitter relative to its neighbours are calculated from the values measured at the fixed receiver using the known positions of the fixed receiver and the transmitters. The timing offsets measured by the mobile terminal can then be used in a calculation based on well-known standard techniques in which the points of intersection of two or more hyperbolic position lines predict the position of the mobile terminal.
The other method (see our EP-B-0303371, WO-A-8901637, U.S. Pat. No. 6,094,168, EP-A-1025453, and WO-A-9921028, the details of which are hereby incorporated by reference and which refer to a system known as Cursor®) makes use of the measurements made by both the fixed receiver and the mobile terminal to calculate the relative time difference between the signals received from each transmitter by both receivers. This results in a calculation based on the intersection of circles centred on the transmitters.
E-OTD methods, as applied to GSM, have been considered for use in wide-band CDMA systems, in particular those within the Universal Mobile Telephone System (UMTS) ‘third generation’ (3G) technologies. Here, E-OTD has been re-named OTDOA, but it suffers from a major problem, the so-called ‘hearability’ problem. In CDMA networks generally, signals are transmitted by the network transmitters all using the same radio-frequency (RF) channel. In UMTS this channel is about 5 MHz wide. The signals from each transmitter are encoded using a unique ‘spreading code’ which allows a mobile terminal to pick out the required signal provided that (a) it knows the spreading code used by that transmitter, and (b) its internal clock is synchronized with the transmitter signals. To assist with the latter, each transmitter also radiates a ‘pilot code’ within the same RF channel whose coding and other characteristics make it easily distinguishable. The mobile terminal first detects and locks on to the pilot signal, receives the spreading code used by that transmitter, and then is able to decode the main transmissions. The hearability problem arises when the mobile terminal is near to a transmitter. E-OTD systems (and therefore OTDOA systems) require the measurements of the time offsets associated with at least three geographically-distinct transmitters, but when the mobile terminal is too close to a transmitter, the signals from the more-distant transmitters are drowned out by the local signals to the extent that their time offsets cannot be measured. One technique, known as ‘Idle Period on the Down Link’ (IP-DL), has been proposed to overcome this problem by which the transmissions from the local transmitter are turned off periodically in a so-called ‘idle period’ during which the signals from the distant transmitters may be received. This has the serious disadvantages that (a) the capacity of the network to carry voice & data traffic is diminished, and (b) it is complicated to install and operate, requiring in one of its forms additional messaging in the network to coordinate the idle periods amongst the transmitters.
An alternative method of countering the hearability problem is described in European patent application number 01306115.5, which provides details of an adaptation of the Cursor® system, especially as described in our U.S. Pat. No. 6,094,168, to CDMA systems in general and particularly to UMTS in such a fashion as to overcome the hearability problem. No idle period is required, and the communications function can therefore operate with full capacity.
The Cursor® system, as described in U.S. Pat. No. 6,094,168, uses two receivers, one fixed and at a known location and the other within the mobile terminal, to receive the signals radiated by each transmitter taken separately. Representations of the received signals are sent back to a computing node where they are compared (generally by cross-correlation) to determine the time offset of receipt of the signals by each receiver. This process is repeated for at least two other geographically-distinct transmitters (transmitting on different RF channels in a GSM system) to obtain the three time offsets required for a successful position computation.
In direct sequence CDMA systems the transmitters use the same RF channel. A direct application of the Cursor® system to CDMA would therefore result in a cross-correlation with many peaks, each corresponding to the alignment of the signals received from a particular one of the transmitters by both receivers. If it were possible to measure the peaks associated with at least the three required transmitters, the system would serve for positioning. However, as illustrated below, the signal to noise ratios (SNRs) associated with more-distant transmitters are often too small, and we have a similar hearability problem as described above.
The following mathematical analysis provides an understanding of the prior art method of countering hearability as described in EP application no. 01306115.5. FIG. 1 shows the geometry of a two-dimensional system in which all the transmitters and the mobile terminal lie in one plane. The positions of transmitters A, B, and C are represented by the vectors a, b, c, all with respect to the same common origin, O. The mobile terminal, R, is at vector position x. Each of the transmitters has incorporated with it a sampling device which samples the signals transmitted by that transmitter and which sends back to a computing device (not shown in FIG. 1) a representation thereof. For simplicity, we make the assumption that the transmitters are synchronized with each other, so that their relative transmission time offsets are known and equal to zero. It is described elsewhere in e.g. U.S. Pat. No. 6,094,168 how the relative transmission time offsets can be measured in unsynchronized networks. Let us suppose that the mobile terminal is nearest to transmitter A, then B, then C. The computing device first performs a cross-correlation between the representation of the signals received (all on the same RF channel) from A, B, and C by R, and the representation of the signals transmitted by A. Since the signals from A, B, and C have orthogonal spreading codes, the cross-correlation results in a single peak whose position represents the time-offset of the receipt of the signals from A by R, together with the clock error, e (in equivalent length dimensions), of the receiver in the mobile terminal. This time offset, ΔtA, is given byvΔtA=|x−a|+ε, where v is the speed of the radio waves, and the vertical bars denote the magnitude of the contained vector quantity. Similarly, for B and C we havevΔtB=|x−b|+ε, vΔtB=|x−c|+ε.  {1}
Having established the time offset of the signals from A, the computing node now subtracts an estimate of the signal received from A by R. The representations of the signals radiated at time t by the transmitters A, B, and C, may be denoted by SA(t), SB(t), and SC(t) respectively. The signal received by the mobile terminal comprises a combination of these. In the absence of multipath, noise and non-linear effects, the representation of the received signals may be denoted by r(t), wherer(t)=αSA(t−ΔtA)+βSB(t−ΔtB)+γSC(t−ΔtC),  {2}and α, β, γ are complex constants representing the path losses to the mobile terminal from the respective transmitters. A software program running in the computing node estimates the magnitude of SA(t), delayed by ΔtA, to subtract from r(t), for example by finding the value of α which minimises the mean square amplitude of the residual r′(t). In the perfect case this would remove the contribution of A altogether, so thatr′(t)=βSB(t−ΔtB)+γSC(t−ΔtC).
The cross-correlation is now carried out between r′(t) and SB(t) to estimate ΔtB, and a further subtraction made to remove the contribution of B from the residual, r″ (t), wherer″(t)=γSC(t−ΔtC),if the subtraction is perfect. Finally, a cross-correlation between r″(t) and SC(t) results in an estimate of ΔtC. Equations {1} can then be solved for x as described in U.S. Pat. No. 6,094,168.
In practice, the signals received by the mobile terminal are corrupted by noise, interference and multipath effects. Furthermore, the representations of the signals may be in a digital format of low resolution. The process of subtraction will not be perfect in these circumstances, but may nevertheless be sufficient to overcome the hearability problem. An example of a prior art system (as proposed in EP application no. 01306115.5) wherein the subtraction is sufficient to overcome the hearability problem will now be described with reference to FIGS. 2 to 7. It is necessary to appreciate fully the prior art in order to understand the advance of the present invention.
FIG. 2 shows a simplified UMTS system consisting of three communications transmitters (Node Bs) 201, 202, 203, each of which has a sampling device 204, 205, 206, a single terminal (user equipment, UE) 207, and a computing device (serving mobile location centre, SMLC) 208. Each Node B has an omni-directional antenna, and is configured to transmit signals typical of network traffic load. Table 1 below indicates the different physical channels in use, together with their power levels and symbol rates. The acronyms appearing in the left-hand column, P-CPICH etc., are those that have been adopted by the industry to represent the channels. Random binary sequences are used to modulate the DPCHs. The three Node Bs use orthogonal primary scrambling codes, in this case numbers 0, 16 and 32 respectively.
TABLE 1Node B channel configurationChannelRelative power Level/dBSymbol rate/Kss−1P-CPICH−1015P-SCH−1015S-SCH−1015P-CCPCH−1015PICH−1515DPCH0Note 1Note 2DPCH1Note 1Note 2DPCH2Note 1Note 2. . .Note 1Note 2. . .Note 1Note 2DPCH63Note 1Note 2DPCH64Note 1Note 2Note 1:DPCH power levels were chosen randomly from −10 dB to −25 dBNote 2:DPCH symbol rates were chosen randomly from 15 to 240 Kss−1
The Node Bs here are tightly synchronised. As already noted above, this is not a requirement in normal practice, but is convenient for the purpose of demonstration.
It will be noted from FIG. 2 that the UE 207 is relatively close to Node B 201 and at greater distances from Node Bs 202 and 203. Thus the signal from Node B 201 is the strongest (0 dB relative to itself) with the signal from Node B 202 weaker at −15 dB and that from Node B 203 weakest of all at −30 dB. The three sampling devices 204, 205, 206 are instructed by the SMLC 208 to record and report the signal transmitted by the associated Node B during the first 256 chips immediately following the start of the next second. These signals are sampled at a rate of 2 samples per chip, with a resolution of 4 bits.
The problem of hearability is highlighted by considering the conventional E-OTD or OTDOA approach to measuring the time offsets of the signals received by the UE 207. A reference copy of the primary scrambling code used on the CPICH by each Node B (i.e. the first 256 chips of each of scrambling codes 0, 16 and 32), is cross-correlated with the signal received by the UE 207 and a search is made for the highest correlation peak. FIG. 3 illustrates a typical result. Note that the signals received by the UE 207 are also sampled at a rate of 2 samples per chip, with a resolution of 4 bits. The resulting cross-correlation profiles show one clearly distinguishable peak 301 in the correlation for scrambling code 0, corresponding to the time offset of the signals from Node B 201. However, the cross-correlation results for the codes 16 and 32 do not yield any clear peaks. This is because the signals received by the UE 207 from Node Bs 202 and 203 are swamped by the relatively strong reception from Node B 201. Were they visible, these peaks should be positioned to the right of the visible peak 301 by 1 and 2 microseconds respectively for the signals from Node Bs 202 and 203 (corresponding to 3.8 and 7.6 chips). The lack of detection of the signals from 202 and 203 means that it is not possible to compute an E-OTD or OTDOA position fix, since at least three independent timings are needed.
The prior art method described in EP 01306115.5 is now illustrated using the same test system. In this case, each sampling device 204, 205, 206 records a section of the signals transmitted by its associated Node B 201, 202, 203 respectively. This section is one symbol in duration and is again sampled at a rate of 2 samples per chip, with a resolution of 4 bits. The UE 207 also records a 256-chip section of the signals it receives, aligned with the first symbol on the CPICH in a particular timeslot, at the same sampling rate and resolution.
At the SMLC 208, the three recordings reported by the three sampling devices 204, 205, 206 are each cross-correlated in turn with the recording made by the UE 207, and the results are shown in FIG. 4. The peaks of the resulting correlation profiles are used to determine the relative levels of the three contributions in the received signal and hence the order in which they are to be subtracted. Once again, the cross-correlation for Node B 201 yields the largest peak 401. Note also that, in contrast with FIG. 3, the cross-correlation for Node B 202 also yields a clear peak 402. This is because the cross-correlation is performed using the total signal transmitted by the Node Bs rather than merely using the CPICH, which represents a fraction of the total transmitted energy in each case.
Having identified the time offset of the signal from Node B 201, the recording of the signal reported by the sampling device 204 is now used to construct an appropriately scaled, delayed and phase-rotated copy of that signal. The results of this process are plotted in FIG. 5. The upper plot shows the real component of the original signal recorded by the UE 207 as a solid curve whilst the dotted curve shows the estimated scaled, delayed and rotated signal. The lower plot shows a similar comparison of the imaginary parts of received and estimated signals. Note that whilst a duration of 256 chips is actually used in the example, the time axis in this Figure has been limited to about 50 chips. The estimated recordings are subtracted from the total UE recording leaving a residual recording.
The recordings from the sampling devices 205 and 206 are now cross-correlated with the residual recording giving the results shown in FIG. 6. Note that in this case, following the removal of the signal from Node B 201, there is a clear correlation peak 601 for the signals from Node B 203 as well as a peak 602 for Node B 202. These peaks are used to estimate the time offsets of the corresponding signals, giving sufficient independent timing measurements (three in this case) to compute a position fix.
If the peak 601 corresponding to the signals from Node B 203 is too weak to be resolved, a further iteration could be undertaken in which the signals from Node B 202 could be subtracted to yield a second residual signal (FIG. 7). There is a clear correlation peak 701 at a delay of approximately 7 chips as expected.
In the prior art method of EP 01306115.5 discussed above, the assumption is made that the signal received at the terminal is a simple sum of the transmitted signals attenuated, phase rotated and delayed by the individual path lengths between transmitter and receiver. In a more complex system in which the transmission channel has incorporated non-linear effects, multi-path and noise, the transmitted signal is further degraded by these effects making the edges of the waveform less clearly defined in time. This process we have called a ‘blurring’ of the signal. When attempting to cancel a blurred signal, the process of subtracting only a simply scaled, delayed and phase rotated copy of the signal recorded by one sampling device from the signal received at the terminal may not remove the contribution from the transmitter associated with the one sampling device accurately enough.